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isolation amplifier with short circuit and overload detection technical data hcpl-788j features ? output voltage directly compatible with a/d converters (0 v to v ref ) ? fast (3 m s) short circuit detection with transient fault rejection 16 15 14 13 12 11 10 9 1 2 3 4 5 6 7 8 fault absval v out v ref v in+ v in- r sense1 short circuit fault isolation boundary a/d converter micro controller hcpl-788j 16 15 14 13 12 11 10 9 1 2 3 4 5 6 7 8 fault absval v out v ref v in+ v in- r sense2 isolation boundary hcpl-788j 16 15 14 13 12 11 10 9 1 2 3 4 5 6 7 8 fault absval v out v ref v in+ v in- r sense3 isolation boundary hcpl-788j m +5 v overload fault + + 3 phase absolute value output v ref v th 3 phase motor low cost three phase current sensing with short circuit and overload detection hewlett-packards isolation amplifier with short circuit and overload detection makes motor phase current sensing compact, affordable and easy-to-implement while satisfying worldwide safety and regulatory requirements. caution: it is advised that normal static precautions be taken in handling and assembly of this component to prevent damage and/ or degradation which may be induced by esd. ? absolute value signal out- put for overload detection ? 1 m v/ c offset change vs. temperature ? so-16 package ? -40 c to +85 c operating temperature range ? 25 kv/ m s isolation transient immunity ? regulatory approvals (pending): ul, csa, vde 0884 (891 vpeak working voltage)
2 symbol description gnd 2 ground input. v dd2 supply voltage input (4.5 v to 5.5 v). fault short circuit fault output. fault changes from a high to low output voltage within 6 m s after v in exceeds the fault detec- tion threshold. fault is an open drain output which allows outputs from all the hcpl-788js in a circuit to be connected together (wired-or) forming a single fault signal for interfacing directly to the micro-controller. absval absolute value of v out output. absval is 0 v when v in =0 and increases toward v ref as v in approaches +256 mv or -256 mv. absval is wired-or able and is used for detecting overloads. v out voltage output. swings from 0 to v ref . the nominal gain is v ref /504 mv. v ref reference voltage input (4.0 v to v dd2 ). this voltage establishes the full scale output ranges and gains of v out and absval. v dd2 supply voltage input (4.5 v to 5.5 v). gnd 2 ground input. pin descriptions symbol description v in+ positive input voltage ( 200 mv recommended). v in- negative input voltage (normally connected to gnd 1 ). c h internal bias node. connections to or between c h and c l other than the re- quired 0.1 m f capacitor shown, are not recommended. c l v dd1 supply voltage input (4.5 v to 5.5 v). v led+ led anode. this pin must be left uncon- nected for guaranteed data sheet perfor- mance. (for optical coupling testing only.) v dd1 supply voltage input (4.5 v to 5.5 v). gnd 1 ground input. description the hcpl-788j isolation amplifier is designed for current sensing in electronic motor drives. in a typical implementation, motor currents flow through an external resistor and the resulting analog voltage drop is sensed by the hcpl-788j. a larger analog output voltage is created on the other side of the hcpl-788js optical isolation barrier. the output voltage is proportional to the motor current and can be connected directly to a single- supply a/d converter. a digital over-range output (fault) and an analog rectified output (absval) are also provided. the wire or-able over-range output (fault) is useful for quick detection of short circuit con- figure 1. current sensing circuit. ditions on any of the motor phases. the wire-or-able rectified output (absval), simplifies measurement of motor load since it performs polyphase rectification. since the common-mode voltage swings several hundred volts in tens of nanoseconds in modern electronic motor drives, the hcpl-788j was designed to ignore very high common-mode transient slew rates (10 kv/ m s). 16 15 14 13 12 11 10 9 1 2 3 4 5 6 7 8 gnd 2 v dd2 fault absval v out v ref v dd2 gnd 2 v in+ v in- c h c l v dd1 v led+ v dd1 gnd 1 r shunt 0.02 w isolated +5 v 4.7 k w 39 w .01 ? 0.1 ? 0.1 ? isolation boundary input current +5 v a/d v ref gnd ? to other phase outputs + 0.1 ? hcpl-788j
3 ordering information specify part number followed by option number (if desired). example hcpl-788j# xxx no option = 16-lead, surface mt. package, 45 per tube. 500 = tape and reel packaging option, 850 per reel. option data sheets available. contact hewlett-packard sales representative, authorized distributor, or visit our web site at www.hp.com/go/isolator. package outline drawings 16-lead surface mount dimensions in inches (millimeters) package characteristics parameter symbol min. typ. max. units test conditions fig. notes input-output momentary v iso 3750 vrms rh < 50%, t = 1 1,2,3 withstand voltage min., t a = 25 c resistance (input-output) r i-o >10 9 w v i-o = 500 v dc 3 capacitance (input-output) c i-o 1.3 pf f = 1 mhz input ic junction-to-case q jci 120 c/w t a = 85 c thermal resistance output ic junction-to-case q jco 100 thermal resistance note: initial and continued variation in the color of the hcpl-788js white mold compound is normal and does not affect device performance or reliability. 240 d t = 115?, 0.3?/sec 0 d t = 100?, 1.5?/sec d t = 145?, 1?/sec time ?minutes temperature ?? 220 200 180 160 140 120 100 80 60 40 20 0 260 123 456789101112 (note: use of non-chlorine activated fluxes is recommended.) maximum solder reflow temperature profile 9 0.295 ?0.010 (7.493 ?0.254) 10 11 12 13 14 15 16 8 7 6 5 4 3 2 1 0.018 (0.457) 0.138 ?0.005 (3.505 ?0.127) 9 0.406 ?0.10 (10.312 ?0.254) 0.408 ?0.010 (10.160 ?0.254) 0.025 min. 0.008 ?0.003 (0.203 ?0.076) standoff 0.345 ?0.010 (8.986 ?0.254) 0? 0.018 (0.457) 0.050 (1.270) all leads to be coplanar ?0.002 hp 788j yyww type number date code
4 regulatory information the hcpl-788j is pending approval by the following organizations: figure 2. dependence of safety- limiting values on temperature. vde 0884 insulation characteristics* description symbol characteristic unit installation classification per din vde 0110/1.89, table 1 for rated mains voltage 300 v rms i-iv for rated mains voltage 300 v rms i-iii for rated mains voltage 600 v rms i-ii climatic classification 55/85/21 pollution degree (din vde 0110/1.89) 2 maximum working insulation voltage v iorm 891 v peak input to output test voltage, method b** v iorm x 1.875 = v pr , 100% production test with v pr 1670 v peak tm = 1 sec, partial discharge < 5 pc input to output test voltage, method a** v iorm x 1.5 = v pr , type and sample test, t m = 60 sec, v pr 1336 v peak partial discharge < 5 pc highest allowable overvoltage v iotm 6000 v peak (transient overvoltage t ini = 10 sec) safety-limiting values maximum values allowed in the event of a failure, also see figure 2. case temperature t s 175 c input power p s1, input 400 mw output power p s1, output 600 mw insulation resistance at t si , v io = 500 v r s >10 9 w * isolation characteristics are guaranteed only within the safety maximum ratings which must be ensured by protective circuits within the application. surface mount classification is class a in accordance with cecc00802. ** refer to the optocoupler section of the isolation and control components designers catalog, under product safety regulations section, (vde-0884) for a detailed description of method a and method b partial discharge test profiles. vde approved under vde0884/06.92 with v iorm = 891 vpeak. ul recognized under ul 1577, com- ponent recognition program, file e55361. csa approved under csa component acceptance notice #5, file ca 88324. p s ?power ?mw 0 0 t s ?case temperature ?? 200 25 800 50 75 100 200 150 175 p si ?output p si ?input 125 400 600
5 insulation and safety related specifications parameter symbol min. max. conditions minimum external air gap l(101) 8.3 mm measured from input terminals to output (clearance) terminals, shortest distance through air. minimum external tracking l(102) 8.3 mm measured from input terminals to output (creepage) terminals, shortest distance path along body. minimum internal plastic gap 0.5 mm through insulation distance conductor to (internal clearance) conductor, usually the straight line distance thickness between the emitter and detector. tracking resistance cti >175 volts din iec 112/vde 0303 part 1 (comparative tracking index) isolation group iiia material group (din vde 0110, 1/89, table 1) recommended operating conditions parameter symbol min. max. units note ambient operating temperature t a -40 85 c supply voltages v dd1 , v dd2 4.5 5.5 v input voltage (accurate and linear) v in+ , v in- -200 200 mv input voltage (functional) v in+ , v in- -2 2 v reference input voltage v ref 4.0 v dd2 fault output current i fault 4ma absolute maximum ratings parameter symbol min. max. units note storage temperature t s -55 125 c operating temperature t a -40 100 supply voltages v dd1 , v dd2 0.0 5.5 v 4 steady-state input voltage v in+ , v in- -2.0 v dd1 + 0.5 2 second transient input voltage -6.0 output voltage v out -0.5 v dd2 + 0.5 absolute value output voltage absval -0.5 v dd2 + 0.5 reference input voltage v ref 0v dd2 + 0.5 v 5 reference input current i ref 20 ma output current i vout 20 absolute value current i absval 20 fault output current i fault 20 input ic power dissipation p i 200 mw output ic power dissipation p o 200 solder reflow temperature profile see package outline drawings section
6 dc electrical specifications unless otherwise noted, all typicals and figures are at the nominal operating conditions of v in+ = 0, v in- = 0 v, v ref = 4.0 v, v dd1 = v dd2 = 5 v and t a = 25 c; all minimum/maximum specifications are within the recommended operating conditions. test parameter symbol min. typ. max. units conditions fig. note input offset v os -3 0 3 mv v in+ = 0 v 3, 4, 6 voltage 5 magnitude of input | d v os / d t a |110 m v/ c7 offset change vs. temperature v out gain g v ref /504 mv - 5% v ref /504 mv v ref /504 mv + 5% v/v |v in+ |<200 mv 6,7, magnitude of v out | d g/ d t a | 50 300 ppm/ 8,9 8 gain change vs. c temperature v out 200 mv nl 200 0.06 0.4 % nonlinearity maximum input |v in+ | max 256 mv voltage before v out clipping fault detection |v thf | 230 256 280 10 9 threshold fault low v olf 350 800 i ol = 4 ma output voltage fault high i ohf 0.2 15 m av fault = v dd2 output current absval output e abs 0.6 2 % of 11 10 error full scale output input supply i dd1 10.7 20 ma current output supply i dd2 10.4 20 current reference voltage i vref 0.26 1 input current input current i in+ -350 na v in+ = 0 v input resistance r in 800 k w v out output r out 0.2 w resistance absval output r abs 0.3 resistance input dc common- cmrr in 85 db 11 mode rejection ratio
7 ac electrical specifications unless otherwise noted, all typicals and figures are at the nominal operating conditions of v in+ = 0, v in- = 0 v, v ref = 4.0 v, v dd1 = v dd2 = 5 v and t a = 25 c; all minimum/maximum specifications are within the recommended operating conditions. parameter symbol min. typ. max. units test conditions fig. note v out bandwidth (-3db) bw 20 30 khz v in+ = 200 mv pk-pk 12, sine wave. 20 v out noise n out 2.2 4 mvrms v in+ = 0 v 20 12 v in to v out signal delay t dsig 920 m sv in+ = 50 mv to 14, 13 (50 - 50%) 200 mv step. 20 v out rise/fall time t rfsig 10 25 (10C90%) absval signal delay t dabs 920 absval rise/fall time t rfabs 10 25 (10C90%) fault detection delay t fhl 36 v in+ = 0 mv to 15, 14 500 mv step. 20 fault release delay t flh 10 20 v in+ = 500 mv to 16, 15 0 mv step. 20 transient fault rejection t reject 12 v in+ = 0 mv to 17, 16 500 mv pulse. 20 common mode transient cmti 10 25 kv/ m s for v out , fault, and 17 immunity absval outputs. common-mode rejection cmrr >140 db 18 ratio at 60 hz
8 notes: 0 1. in accordance with ul1577, each optocoupler is proof tested by applying an insulation test voltage 3 4200 vrms for 1 second (leakage detection current limit, i i-o 5 m a). this test is performed before the 100% production test for partial discharge (method b) shown in vde 0884 insulation characteristic table, if applicable. 0 2. the input-output momentary withstand voltage is a dielectric voltage rating that should not be interpreted as an input-output continuous voltage rating. for the continuous voltage rating refer to your equipment level safety specification or vde0884 insulation characteristics table. 0 3. device considered a two terminal device: pins 1-8 shorted together and pins 9-16 shorted together. 0 4. v dd1 must be applied to both pins 5 and 7. v dd2 must be applied to both pins 10 and 15. 0 5. if v ref exceeds v dd2 (due to power-up sequence, for example), the current into pin 11 (i ref ) should be limited to 20 ma or less. 0 6. input offset voltage is defined as the dc input voltage required to obtain an output voltage (at pin 12) of v ref /2. 0 7. this is the absolute value of input offset change vs. temperature. 0 8. this is the absolute value of v out gain change vs. temperature. 0 9. |v in+ | must exceed this amount in order for the fault output to be activated. 10. absval is derived from v out (which has the gain and offset tolerances stated earlier). absval is 0 v when v in = 0 v and increases toward v ref as v in approaches +256 mv or -256 mv. e abs is the difference between the actual absval output and what absval should be, given the value of v out . e abs is expressed in terms of percent of full scale and is defined as |absval - 2 x | v out - v ref / 2| | x 100. v ref 11. cmrr in is defined as the ratio of the gain for differential inputs applied between pins 1 and 2 to the gain for common mode inputs applied to both pins 1 and 2 with respect to pin 8. 12. the signal-to-noise ratio of the hcpl-788j can be improved with the addition of an external low pass filter to the output. see frequently asked question #4.2 in the applications information section at the end of this data sheet. 13. as measured from 50% of v in to 50% of v out . 14. this is the amount of time from when the fault detection threshold (230 mv v thf 280 mv) is exceeded to when the fault output goes low. 15. this is the amount of time for the fault output to return to a high state once the fault detection threshold (230 mv v thf 280 mv) is no longer exceeded. 16. input pulses shorter than the fault rejection pulse width (t reject ), will not activate the fault (pin 14) output. see frequently asked question #2.3 in the applications information section at the end of this data sheet for additional detail on how to avoid false tripping of the fault output due to cable capacitance charging transients. 17. cmti is also known as common mode rejection or isolation mode rejection. it is tested by applying an exponentially rising/falling voltage step on pin 8 (gnd1) with respect to pin 9 (gnd2). the rise time of the test waveform is set to approximately 50 ns. the amplitude of the step is adjusted until v out (pin 12) exhibits more than 100 mv deviation from the average output voltage for more than 1 m s. the hcpl-788j will continue to function if more than 10 kv/ m s common mode slopes are applied, as long as the break- down voltage limitations are observed. [the hcpl-788j still functions with common mode slopes above 10 kv/ m s, but output noise may increase to as much as 600 mv peak to peak.] 18. cmrr is defined as the ratio of differential signal gain (signal applied differentially between pins 1 and 2) to the common mode gain (input pins tied to pin 8 and the signal applied between the input and the output of the isolation amplifier) at 60 hz, expressed in db.
9 figure 3. input offset voltage change vs. temperature. figure 4. input offset voltage change vs. v dd1 . figure 5. input offset voltage change vs. v dd2 . figure 6. v out vs. v in . figure 7. gain change vs. temperature. figure 8. gain change vs. v dd1 . d v os offset change ?? 4.5 -800 input supply voltage ?v dd1 ?v 800 4.75 5.0 5.25 5.5 0 600 400 200 -200 -400 -600 d v os offset change ?? 4.5 -800 output supply voltage ?v dd2 ?v 800 4.75 5.0 5.25 5.5 0 600 400 200 -200 -400 -600 v out ?output voltage ?v -300 0 input voltage ?v in ?mv 4.0 -200 0 100 300 2.0 3.5 3.0 2.5 1.5 1.0 0.5 -100 200 d gain change-% -40 -2.0 temperature ?? -20 2.0 020 -1.0 60 80 typical worst case 40 0 1.0 1.5 0.5 -0.5 -1.5 d gain change-% 4.5 -2.0 input supply voltage ?v dd1 ?v 2.0 4.75 5.0 5.25 5.5 0 1.5 1.0 0.5 -0.5 -1.0 -1.5 figure 9. gain change vs. v dd2 . figure 10. fault output voltage vs. v in . figure 11. absval output voltage vs. v in . d gain change-% 4.5 -2.0 output supply voltage ?v dd2 ?v 2.0 4.75 5.0 5.25 5.5 0 1.5 1.0 0.5 -0.5 -1.0 -1.5 fault output voltage ?faultbar ?v -300 0 input voltage ?v in ?mv 5.0 -200 0 100 300 2.0 3.5 3.0 2.5 1.5 1.0 0.5 -100 200 4.0 4.5 absval ?absolute value output ?v -300 0 input voltage ?v in ?mv 4.0 -200 0 100 300 2.0 3.5 3.0 2.5 1.5 1.0 0.5 -100 200 input offset change - d v os - uv -40 -800.0 temperature ?deg c -20 800.0 020 -400.0 60 80 typical max 40 0 400.0 600.0 200.0 -200.0 -600.0
10 figure 15. fault detection, 0 to 300 mv input, at v ref = 5 v. figure 12. bandwidth vs. temperature. figure 13. fault detection delay vs. temperature. figure 14. step response, 0 to 200 mv input, at v ref = 5 v. figure 16. fault release, 300 to 0 mv input, at v ref = 5 v. figure 17. fault rejecting a 1 m s, 0 to 2 v to 0 input. rejection is independent of amplitude. figure 18. detection of 6 m s fault 0 to 2 v to 0 input, at v ref = 5 v. figure 19. sine response 400 mv pk to pk 4 khz input, at v ref = 5 v. bandwidth ?khz -40 25 temperature ?? -20 35 020 6080 40 30 34 33 32 31 29 28 27 26 fault detection delay ?? -40 2.5 temperature ?? -20 3.5 020 6080 40 3.0 2.75 3.25 0 v 2.5 v 5 v 0 v 2.5 v 5 v 0 v 2.5 v 5 v -300 mv 0 mv 300 mv 5.00 ?/div v in 300 mv/d v out (pin 12) 2.5 v/d fault (pin 14) 2.5 v/d absval (pin 13) 2.5 v/d 0 v 2.5 v 5 v 0 v 2.5 v 5 v 0 v 2.5 v 5 v -300 mv 0 mv 300 mv 5.00 ?/div v in 300 mv/d v out (pin 12) 2.5 v/d absval (pin 13) 2.5 v/d fault (pin 14) 2.5 v/d 0 v 2.5 v 5.0 v 0 v 2.5 v 5.0 v 0 v 2.5 v 5.0 v -2 v 0 mv 2 v 5.00 ?/div v in 2.0 v/d v out (pin 12) 2.5 v/d absval (pin 13) 2.5 v/d fault (pin 14) 2.5 v/d 0 v 2.5 v 5 v 0 v 2.5 v 5 v 0 v 2.5 v 5 v -300 mv 0 mv 300 mv 100 ?/div v in 300 mv/d v out (pin 12) 2.5 v/d absval (pin 13) 2.5 v/d fault (pin 14) 2.5 v/d 0 v 2.5 v 5 v 0 v 2.5 v 5 v 0 v 2.5 v 5 v -300 mv 0 mv 300 mv 5.00 ?/div v in 300 mv/d v out (pin 12) 2.5 v/d absval (pin 13) 2.5 v/d fault (pin 14) 2.5 v/d 5.00 ?/div 0 v 2.5 v 5 v 0 v 2.5 v 5 v 0 v 2.5 v 5 v -300 mv 0 mv 300 mv fault (pin 14) 2.5 v/d v in 300 mv/d v out (pin 12) 2.5 v/d absval (pin 13) 2.5 v/d
11 figure 20. ac test circuit. 14 13 12 11 10,15 9, 16 1 3 4 6 5, 7 2, 8 fault absval v out v dd2 v in+ v dd1 4.7 k w 50 w 0.01 ? 0.1 ? 0.1 ? hcpl-788j v ref 10 w 0.1 ? 0.1 ? 5 v figure 21. internal block diagram. 1 v ref v out fault absval v dd2 v dd2 gnd 2 gnd 2 v in+ v in- c h c l v led+ v dd1 v dd1 gnd 1 2 3 4 6 5 7 8 11 12 13 14 15 10 9 16 sd modulator 256 mv reference fault detect encoder rectifier decoder d/a lpf hcpl-788j
12 figure 24. absval with 1 phase. figure 23. absval with 2 phases, wired-ored together. figure 22. absval with 3 phases, wired-ored together. absval ?v 0 0 time ?seconds 4.0 0.01 0.02 0.03 0.04 2.0 3.0 1.0 absval ?v 0 0 time ?seconds 4.0 0.01 0.02 0.03 0.04 2.0 3.0 1.0 absval ?v 0 0 time ?seconds 4.0 0.01 0.02 0.03 0.04 2.0 3.0 1.0 applications information production description figure 21 shows the internal block diagram of the hcpl-788j. the analog input (v in ) is con- verted to a digital signal using a sigma-delta ( ? - d ) analog to digital (a/d) converter. this a/d samples the input 6 million times per second and generates a high speed 1-bit output representing the input very accurately. this 1 bit data stream is transmitted via a light emitting diode (led) over the optical barrier after encoding. the detector converts the optical signal back to a bit stream. this bit stream is decoded and drives a 1 bit digital to analog (d/a) con- verter. finally a low pass filter and output buffer drive the output signal (v out ) which linearly rep- resents the analog input. the output signal full-scale range is determined by the external reference voltage (v ref ). by sharing this reference voltage (which can be the supply voltage), the full-scale range of the hcpl-788j can precisely match the full-scale range of an external a/d converter. in addition, the hcpl-788j compares the analog input (v in ) to both the negative and positive full-scale values. if the input exceeds the full-scale range, the short-circuit fault output (fault) is activated quickly. this feature operates indepen- dently of the ? - d a/d converter in order to provide the high- speed response (typically 3 m s) needed to protect power tran- sistors. the fault output is wire or-able so that a short circuit on any one motor phase can be detected using only one signal. one other output is provided the rectified output (absval). this output is also wire or-able. the motor phase having the highest instantaneous rectified output pulls the common output high. when three sinusoidal motor phases are combined, the rectified output (absval) is essentially a dc signal represent- ing the rms motor current. this single dc signal and a threshold comparator can indicate motor overload conditions before dam- age to the motor or drive occur. figure 22 shows the absval output when 3 hcpl-788js are used to monitor a sinusoidal 60 hz current. figures 23 and 24 show the absval output when only 2 or 1 of the 3 phases are monitored, respectively. the hcpl-788js other main function is to provide galvanic isolation between the analog input and the analog output. an internal voltage reference determines the full-scale analog input range of the modulator (approximately 256 mv); an input range of 200 mv is recommended to achieve optimal performance.
13 analog interfacing power supplies and bypassing the recommended supply con- nections are shown in figure 26. a floating power supply (which in many applications could be the same supply that is used to drive the high-side power transistor) is regulated to 5 v using a simple zener diode (d1); the value of resistor r4 should be chosen to supply sufficient current from figure 25. recommended applications circuit. figure 26. recommended supply and sense resistor connections. the existing floating supply. the voltage from the current sensing resistor (rsense) is applied to the input of the hcpl-788j through an rc anti-aliasing filter (r2 and c2). although the application circuit is relatively simple, a few recommendations should be followed to ensure optimal performance. the power supply for the hcpl-788j is most often obtained from the same supply used to power the power transistor gate drive circuit. if a dedicated supply is required, in many cases it is possible to add an additional winding on an existing trans- former. otherwise, some sort of simple isolated supply can be used, such as a line powered 16 15 14 13 12 11 10 9 1 2 3 4 5 6 7 8 gnd 2 v dd2 fault absval v out v ref v dd2 gnd 2 v in+ v in- c h c l v dd1 v led1+ v dd1 gnd 1 r shunt 0.02 w isolated +5 v r3 4.7 k w r2 39 w .01 ? c3 0.1 ? c1 0.1 ? hcpl-788j input current +5 v a/d v ref gnd ? to other phase outputs + r1 c2 c6 0.1 ? c4 c8 c7 c5 c5 = c7 = c8 = 470 pf c4 = 0.1 ? 16 15 14 13 12 11 10 9 5 1 2 8 7 3 4 6 gnd 2 v dd2 fault absval v out v ref v dd2 gnd 2 v dd1 v in+ v in- gnd 1 v dd1 c h c l v led+ r2 39 w + r4 c1 0.1 ? c2 0.01 ? floating power supply hv+ + + r1 r sense hv- motor d1 5.1 v gate drive circuit hcpl-788j
14 transformer or a high-frequency dc-dc converter. an inexpensive 78l05 three- terminal regulator can also be used to reduce the floating supply voltage to 5 v. to help attenuate high-frequency power supply noise or ripple, a resistor or inductor can be used in series with the input of the regulator to form a low-pass filter with the regulators input bypass capacitor. as shown in figure 25, 0.1 m f bypass capacitors (c1, c3, c4, and c6) should be located as close as possible to the pins of the hcpl-788j. the bypass capacitors are required because of the high-speed digital nature of the signals inside the hcpl-788j. a 0.01 m f bypass capacitor (c2) is also recommended at the input due to the switched-capacitor nature of the input circuit. the input bypass capacitor also forms part of the anti-aliasing filter, which is recommended to prevent high-frequency noise from aliasing down to lower frequencies and interfering with the input signal. the input filter also performs an important reliability function it reduces transient spikes from esd events flowing through the current sensing resistor. pc board layout the design of the printed circuit board (pcb) should follow good layout practices, such as keeping bypass capacitors close to the supply pins, keeping output signals away from input signals, the use of ground and power planes, etc. in addition, the layout of the pcb can also affect the isolation transient immunity (cmti) of the hcpl-788j, due primarily to stray capacitive coupling between the input and the output circuits. to obtain optimal cmti performance, the layout of the pc board should minimize any stray coupling by maintaining the maximum possible distance between the input and output sides of the circuit and ensuring that any ground or power plane on the pc board does not pass directly below or extend much wider than the body of the hcpl-788j. figure 27. example printed circuit board layout. top layer bottom layer
15 current sensing resistors the current sensing resistor should have low resistance (to minimize power dissipation), low inductance (to minimize di/dt induced voltage spikes which could adversely affect operation), and reasonable tolerance (to maintain overall circuit accuracy). choosing a particular value for the resistor is usually a compro- mise between minimizing power dissipation and maximizing ac- curacy. smaller sense resistance decreases power dissipation, while larger sense resistance can improve circuit accuracy by utilizing the full input range of the hcpl-788j. the first step in selecting a sense resistor is determining how much current the resistor will be sens- ing. the graph in figure 28 shows the rms current in each phase of a three-phase induction motor as a function of average motor output power (in horse- power, hp) and motor drive supply voltage. the maximum value of the sense resistor is determined by the current being measured and the maximum recommended input voltage of the isolation amplifier. the maxi- mum sense resistance can be calculated by taking the maxi- mum recommended input voltage and dividing by the peak current that the sense resistor should see during normal operation. for example, if a motor will have a maximum rms current of 10 a and can experience up to 50% overloads during normal operation, then the peak current is 21.1 a (=10 x 1.414 x 1.5). assuming a maximum input voltage of 200 mv, the maximum value of sense resistance in this case would be about 10 m w . the maximum average power dissipation in the sense resistor can also be easily calculated by multiplying the sense resistance times the square of the maximum rms current, which is about 1 w in the previous example. if the power dissipation in the sense resistor is too high, the resistance can be decreased below the maximum value to decrease power dissipation. the minimum value of the sense resistor is limited by precision and accuracy requirements of the design. as the resistance value is reduced, the output voltage across the resistor is also reduced, which means that the offset and noise, which are fixed, become a larger percentage of the signal amplitude. the selected value of the sense resistor will fall somewhere between the minimum and maximum values, depending on the particular requirements of a specific design. when sensing currents large enough to cause significant heating of the sense resistor, the temperature coefficient (tempco) of the resistor can introduce nonlinearity due to the signal dependent temperature rise of the resistor. the effect increases as the resistor-to-ambient thermal resistance increases. this effect can be minimized by reducing the thermal resistance of the current sensing resistor or by using a resistor with a lower tempco. lowering the thermal resistance can be accomplished by reposi- tioning the current sensing resistor on the pc board, by using larger pc board traces to carry away more heat, or by using a heat sink. for a two-terminal current sensing resistor, as the value of resistance decreases, the resistance of the leads become a significant per- centage of the total resistance. this has two primary effects on resistor accuracy. first, the effective resistance of the sense resistor can become dependent on factors such as how long the leads are, how they are bent, how far they are inserted into the board, and how far solder wicks up the leads during assembly (these issues will be discussed in more detail shortly). second, the leads are typically made from a material, such as copper, which has a much higher tempco than the material from which the resis- tive element itself is made, result- ing in a higher tempco overall. both of these effects are eliminated when a four-terminal current sensing resistor is used. a four- terminal resistor has two additional terminals that are kelvin-connected directly across the resistive element itself; these two terminals are used to monitor the voltage across the resistive element while the other two terminals are used to carry the load current. because of the kelvin connection, any voltage drops across the leads carrying figure 28. motor output horsepower vs. motor phase current and supply voltage. motor output power ?horsepower 0 0 motor phase current ?a (rms) 40 5202535 20 35 30 25 15 10 5 10 15 30 440 380 220 120
16 the load current should have no impact on the measured voltage. when laying out a pc board for the current sensing resistors, a couple of points should be kept in mind. the kelvin connections to the resistor should be brought together under the body of the resistor and then run very close to each other to the input of the hcpl-788j; this minimizes the loop area of the connection and reduces the possibility of stray magnetic fields from interfering with the measured signal. if the sense resistor is not located on the same pc board as the hcpl-788j circuit, a tightly twisted pair of wires can accomplish the same thing. also, multiple layers of the pc board can be used to increase current carrying capacity. numerous plated-through vias should surround each non-kelvin terminal of the sense resistor to help distribute the current between the layers of the pc board. the pc board should use 2 or 4 oz. copper for the layers, resulting in a current carrying capacity in excess of 20 a. making the current carrying traces on the pc board fairly large can also improve the sense resistors power dissipation capability by acting as a heat sink. liberal use of vias where the load current enters and exits the pc board is also recommended. sense resistor connections the recommended method for connecting the hcpl-788j to the current sensing resistor is shown in figure 26. v in+ (pin 1 of the hcpl-788j) is connected to the positive terminal of the sense resistor, while v in- (pin 2) is shorted to gnd 1 (pin 8), with the power-supply return path functioning as the sense line to the negative terminal of the current sense resistor. this allows a single pair of wires or pc board traces to connect the hcpl-788j circuit to the sense resistor. by referencing the input circuit to the negative side of the sense resistor, any load current induced noise transients on the resistor are seen as a common- mode signal and will not interfere with the current-sense signal. this is important because the large load currents flowing through the motor drive, along with the parasitic inductances inherent in the wiring of the circuit, can generate both noise spikes and offsets that are rela- tively large compared to the small voltages that are being measured across the current sensing resistor. if the same power supply is used both for the gate drive circuit and for the current sensing circuit, it is very important that the connection from gnd 1 of the hcpl-788j to the sense resistor be the only return path for supply current to the gate drive power supply in order to eliminate potential ground loop problems. the only direct connection between the hcpl-788j circuit and the gate drive circuit should be the positive power supply line. please refer to hewlett-packard applications note 1078 for additional information on using isolation amplifiers.
17 1. the basics historically, motor control current sense designs have required trade-offs between signal accuracy, response time, and the use of discrete components to detect short circuit and overload conditions. the hcpl-788j greatly simplifies current-sense designs by providing an output voltage which can connect directly to an a/d converter as well as integrated short circuit and overload detection (eliminating the need for external circuitry). available in an auto-insertable, so-16 package, the hcpl-788j is smaller than and has better linearity, offset vs. temperature and common mode rejection (cmr) performance than most hall-effect sensors. the v ref input establishes the full scale output range. v ref can be connected to the supply voltage (v dd2 ) or a voltage between 4 v and v dd2 . the nom- inal gain of the hcpl-788j is the output full scale range divided by 504 mv. when 3 phases are wire-ored together, the 3 phase ac currents are combined to form a dc voltage with very little ripple on it. this can be simply filtered and used to monitor the motor load. moderate overload currents which dont trip the fault output can thus be detected easily. frequently asked questions about the hcpl-788j 1.1: why should i use the hcpl-788j for sensing current when hall-effect sensors are available which dont need an isolated supply voltage? 1.2: what is the purpose of the v ref input? 1.3: what is the purpose of the rectified (absval) output on pin 13? 2. sense resistor and input filter although less common than values above 10 w , there are quite a few manufacturers of resistors suitable for measuring currents up to 50 a when combined with the hcpl-788j. example product information may be found at dales web site (http://www.vishay.com/vishay/dale) and isoteks web site (http://www.isotekcorp.com). this is not necessary, but it will work. if you do, be sure to use an rc filter on both pin 1 (v in+ ) and pin 2 (v in- ) to limit the input voltage at both pads. in pwm motor drives there are brief spikes of cur- rent flowing in the wires leading to the motor each time a phase voltage is switched between states. the amp- litude and duration of these current spikes is deter- mined by the slew rate of the power transistors and the wiring impedances. to avoid false tripping of the fault output (pin 14) the hcpl-788j includes a blanking filter. this filter ignores over-range input conditions shorter than 1 m s. for very long motor wires, it may be necessary to increase the time con- stant of the input rc anti-aliasing filter to keep the peak value of the hcpl-788j inputs below 230 mv. for example, a 39 w , 0.047 m f rc filter on pin 1 will ensure that 2 m s wide 500 mv pulses across the sense resistor do not trip the fault output. 2.1: where do i get 10 m w resistors? i have never seen one that low. 2.2: should i connect both inputs across the sense resistor instead of grounding v in- directly to pin 8? 2.3: how can i avoid false tripping of the fault output due to cable capacitance charging transients?
18 this filter prevents damage from input spikes which may go beyond the absolute maximum ratings of the hcpl-788j inputs during esd and other transient events. the filter also prevents aliasing of high fre- quency (above 3 mhz) noise at the sampled input. other rc values are certainly ok, but should be chosen to prevent the input voltage (pin 1) from ex- ceeding 5 v for any conceivable current waveform in the sense resistor. remember to account for in- ductance of the sense resistor since it is possible to momentarily have tens of volts across even a 1 m w resistor if di/dt is quite large. select the sense resistor so that it will have less than 5 v drop when short circuits occur. the ony other requirement is to shut down the drive before the sense resistor is damaged or its solder joints melt. this ensures that the input of the hcpl-788j can not be damaged by sense resistors going open-circuit. 2.4: do i really need an rc filter on the input? what is it for? are other values of r and c okay? 2.5: how do i ensure that the hcpl-788j is not destroyed as a result of short circuit conditions which cause voltage drops across the sense resistor that exceed the ratings of the hcpl-788js inputs? 3. isolation and insulation the momentary (1 minute) withstand voltage is 3500 v rms per ul1577 and csa component acceptance notice #5. these capacitors are to reduce the narrow output spikes caused by high common mode slew rates. if your application does not have rapid common mode voltage changes, these capacitors are not needed. 3.1: how many volts will the hcpl-788j withstand? 3.2: what happens if i dont use the 470 pf output capacitors hp recommends? 4. accuracy for zero input, the output should ideally be 1 / 2 of v ref . the nominal slope of the input/output relation- ship is v ref divided by 0.504 v. offset errors change only the dc input voltage needed to make the output equal to 1 / 2 of v ref . gain errors change only the slope of the input/output relationship. for example, if v ref is 4.0 v, the gain should be 7.937 v/v. for zero input, the output should be 2.000 v. input offset voltage of 3 mv means the output voltage will be 2.000 v 0.003*7.937 or 2.000 23.8 mv when the input is zero. gain tolerance of 5% means that the slope will be 7.937 0.397. over the full range of 3 mv input offset error and 5% gain error, the output voltage will be 2.000 25.0 mv when the input is zero. 4.1: what is the meaning of the offset errors and gain errors in terms of the output?
19 yes. some noise energy exists beyond the 30 khz bandwidth of the hcpl-788j. an external rc low pass filter can be used to improve the signal to noise ratio. for example, a 680 w , 4700 pf rc filter will cut the rms output noise roughly by a factor of 2. this filter reduces the -3db signal bandwidth only by about 10%. in applications needing only a few khz bandwidth even better noise performance can be obtained. the noise spectral density is roughly 400 nv/ ? hz below 15 khz (input referred). as an example, a 2 k hz (680 w , 0.1 m f) rc low pass filter reduces output noise to a typical value of 0.08 mvrms. at present hp does not have a standard product with tighter gain tolerance. a 100 w variable resistor divider can be used to adjust the input voltage at pin 1, if needed. op-amps are used to drive v out (pin 12) and absval (pin 13). these op-amps can swing nearly from rail to rail when there is no load current. the internal v dd2 is about 100 mv below the external v dd2 . in addition, the pullup and pulldown output transistors are not identical in capability. the net result is that the output can typically swing to within 20 mv of gnd 2 and to within 150 mv of v dd2 . when v ref is tied to v dd2 , the output can not reach v ref exactly. this limitation has no effect on gain only on maximum output voltage. the output remains linear and accurate for all inputs between -200 mv and +200 mv. for the maximum possible swing range, separate v ref and v dd2 voltages can be used. since 5.0 v is normally recommended for v dd2 , use of 4.5 v or 4.096 v references for v ref allow the out- puts to swing all the way up to v ref (and down to typically 20 mv). no. the led is used only to transmit a digital pattern. gain is determined by a bandgap voltage reference and the user-provided v ref . hp has accounted for led degradation in the design of the product to ensure long life. ideally gain would be v ref /512 mv, however, due to internal settling characteristics, the average effective value of the internal 256 mv reference is 252 mv. 4.2: can the signal to noise ratio be improved? 4.3: i need 1% tolerance on gain. does hp sell a more precise version? 4.4: the output doesnt go all the way to v ref when the input is above full scale. why not? 4.5: does the gain change if the internal led light output degrades with time? 4.6: why is gain defined as v ref /504 mv, not v ref /512 mv as expected, based on figure 24?
5. power supplies and start-up v out (pin 12) is close to zero volts, absval (pin 13) is close to v ref and fault (pin 14) is in the high (inactive) state when power to the input side is off. in fact, a self test can be performed using this infor- mation. in a motor drive, it is possible to turn off all the power transistors and thus cause all the sense resistor voltages to be zero. in this case, finding v out less than 1 / 4 of v ref , absval more than 3 / 4 of v ref and fault in the high state indicates that power to the input side is not on. about 50 m s after a v dd2 power-up and 100 m s after a v dd1 power-up. 5.1 : what are the output voltages before the input side power supply is turned on? 5.2: how long does the hcpl-788j take to begin working properly after power-up? 6. miscellaneous the inputs have a series resistor for protection against large negative inputs. normal signals are no more than 200 mv in amplitude. such signals do not forward bias any junctions sufficiently to interfere with accurate operation of the switched capacitor input circuit. typically, noticeable ringing and overshoot begins for c load above 0.02 m f. hp recommends keeping the load capacitance under 5000 pf (at pin 12). absval (pin 13) typically exhibits no instability at any load capacitance, but speed of response gradually slows above 470 pf load. yes, with a compromise on offset accuracy. one way to do this is by connecting +2.5 v to pins 10, 11, and 15 and connecting -2.5 v to pins 9 and 16 with 0.1 m f bypass capacitors from +2.5 v to -2.5 v and from -2.5 v to ground. note that fault cannot swing above 2.5 v in this case, so a level shifter may be needed. alternately, a single 5 v supply could be power the hcpl-788j which could drive an op amp configured to subtract 1 / 2 of v ref from v out . 6.1: how does the hcpl-788j measure negative signals with only a +5 v supply? 6.2: what load capacitance can the hcpl-788j drive? 6.3: can i use the hcpl-788j with a bipolar input a/d converter? for technical assistance or the location of your nearest hewlett-packard sales office, distributor or representative call: americas/canada: 1-800-235-0312 or 408-654-8675 far east/australasia: call your local hp sales office. japan: (81 3) 3335-8152 europe: call your local hp sales office. data subject to change. copyright ? 1997 hewlett-packard co. printed in u.s.a. 5966-0001e (7/97)

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